There are many control methods for controlling an output current in switching mode power supplies (SMPS). BCD Semiconductor provides a primary side controller AP3708 (Preliminary Datasheet, Rev.1.0, 2008.09, http://www.bcdsemi.com) for controlling an output voltage and an output current. FIG. 1A schematically shows a switching mode power supply incorporating the controller AP3708.
In FIG. 1A, an input signal is applied to a first terminal of a primary winding Np. A second terminal of the primary winding Np is coupled to a transistor Q1, which is controlled to be on and off by the controller AP3708. When the transistor Q1 is turned on, a primary current Ip flows through the primary winding Np, which starts building up a magnetic energy. A secondary winding Ns coupled with a secondary diode D1 and a capacitor Co is magnetically coupled to the primary winding Np, wherein the capacitor Co has a relatively high capacity to stabilize the variation of a voltage Vo supplying to a load. The polarity (winding sense) of the secondary winding Ns is configured such that the magnetic field produced by the primary current Ip (when the transistor Q1 is turned on) induces a voltage that reverse biases the secondary diode D1. When the transistor Q1 is turned off, the sign of the time derivative of the magnetic field is reversed and a current Is is induced in the secondary winding Ns. A part of the current Is charges the capacitor Co and the rest is supplied to the load Ro. The capacitor Co maintains an output current lo flowing in the load by partly discharging while the secondary current Is stops flowing. This is the case when the energy in the magnetic field has been completely discharged.
In order to maintain a constant current at the load Ro, an auxiliary winding NAUX may be used. In this example, the auxiliary winding NAUX has the same polarity as that of the secondary winding Ns, and is coupled with a auxiliary diode D2 and a capacitor CAUX The auxiliary diode D2 coupled to the auxiliary winding NAUX is also reverse biased when the transistor Q1 is turned on; and an induced current is flowing through the auxiliary winding NAUX when the transistor Q1 is turned off. The induced current flowing through the auxiliary winding NAUX could be used as an indicator for the controller AP3708 to control switching the transistor Q1 to obtain a constant load current. However, the induced current could not be easily monitored. An alternative solution is to monitor a voltage waveform at the node 101 of the auxiliary winding NAUX. A resistor R1 is coupled to the node 101 to pick up the voltage waveform at the node 101, which may be further attenuated with a resistor R2 to form a feedback signal VFB at the node 102. In one embodiment, the feedback signal VFB may be an image of the voltage Vs across the secondary winding Ns, i.e., VFB is in the first order linearly proportional to the voltage at node 103 adjusted by a turns ratio and the voltage divider ratio.
While the transistor Q1 is turned on, the primary current Ip flows through a current sensing resistor Rcs, which produces a voltage Vcs to the controller AP3708. In one embodiment, the transistor Q1 is turned on when Vcs is below a predetermined value, and the transistor Q1 is turned off when Vcs is above the predetermined value.
FIG. 1B schematically shows a block diagram of the controller AP3708. In FIG. 1B, the constant current control circuit of the controller AP3708 comprises: a first comparator 201, a Tons (Tons represents the conduction time of the secondary diode D1) detector 202, a first current source 203, a second current source 204, a first switch 205, a second switch 206, a capacitor 207, a second comparator 208 and a flip-flop 209.
The feedback signal VFB is compared to a reference signal, e.g., 0.1 Volts by the comparator 201. When the feedback signal VFB is larger than 0.1 Volts, the comparator 201 generates a logical low signal. Otherwise, the comparator generates a logical high signal.
The output of the comparator 201 is coupled to the Tons detector 202. Based on the output of the comparator 201 and a signal pfm which controls the on and off of the transistor Q1, the Tons detector 202 generates a conduction time interval signal Tons indicative of the conduction of the secondary diode D1.
In FIG. 1B, the conduction time interval signal Tons and a non-conduction time interval signal Toffs, which is opposite from the signal Tons, have a ratio of 4/3, i.e., Tons has a relative interval time of 4 whereas Toffs has a relative interval time of 3.
The capacitor 207 is charged by the first current source 203 when the first switch 205 is turned on and is discharged by the second current source 204 when the second switch 206 is turned on. The first switch 205 and the second switch 206 is controlled by the signal Tons. An inverter 210 allows the second switch 206 to be off (i.e., open) when the first switch 205 is turned on (i.e., closed), and to be on when the first switch 205 is turned off. The ratio of the first current source 203 and the second current source 204 is fixed to 4/3.
When the controller AP3708 works under constant current mode, Tons/Toffs=4/3, wherein Toffs=T−Tons, and T represents a switching cycle time period of the transistor Q1. Ideally, when the controller AP3708 works under constant current mode, the output current is:
                    Io        =                              1            2                    ×          n          ×          Ipk          ×                      Tons            T                                              (        1        )            
Wherein Ipk represents the peak current of the primary winding Np, and n represents the turn ratio of the primary winding Np and the secondary winding Ns. So n×Ipk represents the peak current of the current flowing through the secondary winding Ns.
In a given system, n and Ipk are fixed. The output current Io could be constant by fixing the ratio of the cycle time T and the conduction time interval signal Tons. So detecting and controlling the conduction time interval signal Tons is the key to the constant current control. The common way to generate the signal Tons is to detect the zero cross of the voltage VAUX across the auxiliary winding NAUX. The controller AP3708 gets the conduction time interval signal Tons by comparing the feedback signal VFB to a reference signal 0.1V.
In real world application, when the secondary diode D1 is turned off, the voltage VAUX across the auxiliary winding NAUX will cross zero after a demagnetizing oscillation, which means that when the voltage Vs across the secondary winding Ns reduces to zero, the feedback signal VFB is still larger than zero because of the demagnetizing oscillation of the auxiliary winding NAUX. FIG. 2 shows typical waveforms of the feedback signal VFB and the current Is flowing through the secondary winding when the transistor Q1 is turned on and off. At time t0, the transistor Q1 is turned off, and the energy of the magnetic field in the primary winding Np is transferred to the respective secondary and auxiliary windings Ns and NAUX. The secondary diode D1 and the auxiliary diode D2 are conducted. Accordingly, a peak current Ipk flows through the secondary diode D1, and the feedback signal VFB shows some ringing or oscillations before settling down to an average value while the current Is is flowing. At time t1, the energy of the magnetic field stored in the primary winding Np is completely discharged, there are no currents flowing in the primary, secondary and auxiliary windings. As the current Is flowing through the secondary winding Ns drops to zero, the feedback signal VFB shows a series of undershoots and overshoots with damping magnitudes around the ground potential. The undershoots and overshoots have a damped sinusoid ringing waveform with an approximately constant frequency, and the DC level of the ringing waveform is zero. At time t4, the transistor Q1 is turned on, VFB is going negative, i.e., the secondary diode D1 and the diode coupled to the auxiliary winding NAUX are reverse-biased. The current Ip starts to flow through the primary winding Np, which stores the energy of the magnetic field and releases it again to the secondary Ns and the auxiliary windings NAUX at time t0′ when transistor Q1 is turned off. The frequency f of the sinusoidal ringing waveform is determined by the inductance Lp of the primary winding Np, its parasitic capacitance, and the other capacitances related to the PCB layout of the power supply. The frequency f of the sinusoidal ringing waveform is calculated by the following expression:
                    f        =                  1                      2            ⁢            π            ⁢                                          Lp                ×                Ctot                                                                        (        2        )            
Wherein Ctot represents the total capacitance. So a period T of the sinusoidal ringing waveform is:T=2π√{square root over (Lp×Ctot)}  (3)
As shown in FIG. 2, while the current Is drops to zero at time t1, the feedback signal VFB is not yet zero due to the ringing waveform. In some examples, the feedback signal VFB goes to zero after approximately a quarter of one cycle period of the sinusoidal. When a PWM (pulse width modulation) or PFM (pulse frequency modulation) controller utilizes the zero crossing of the feedback signal VFB as an indicator for the conduction time of the diode D1, as shown in FIG. 2, an error ΔT is likely to be included, which corresponds to about a quarter of one cycle period of the sinusoidal waveform.
FIG. 3 schematically shows the section of the constant current control circuit of the controller AP3708. The value of the first current source 203 is marked as 4I and the value of the second current source 204 is marked as 3I for illustration purpose. Ideally, when the controller AP3708 works under constant current mode, Tons/Toffs=4/3. If the error ΔT is considered, we could get the expression:3I×(Tons+ΔT)=4I×(Toffs−ΔT)   (4)
Substitution of Eq. (4) into Eq. (1) and the solution for the signal Tons yields:
                    Io        =                                            2              7                        ×            n            ×            Ipk            ×                          Tons                              Tons                +                                  Δ                  ⁢                                                                          ⁢                  T                                                              =                                    2              ×              n              ×              Lp              ×                              Ipk                2                                                    7              ⁢                              (                                                      Lp                    ×                    Ipk                                    +                                      n                    ×                    Δ                    ⁢                                                                                  ⁢                    T                    ×                    Vo                                                  )                                                                        (        5        )            
Wherein Vo represents the output voltage of the secondary winding. As seen from FIG. 2, the real zero cross of the voltage VFB is happened at t2 because of the demagnetizing oscillation. In Eq. (5), nΔTVo is an error in the output current lo caused by the error ΔT in the time interval of the signal Tons. The output voltage Vo is varied with the load in constant current control mode, thus the output current lo could not be constant because of the varying error nΔTVo.
To solve the above problem, there are several methods. For example, technology in the patent application US 2010/0238689 suggests compensating the time error to the conduction time interval signal Tons. It may solve the problem, but complicated circuits and a substantially similar capacitor set are needed to be configured. Also, a pair of substantially similar capacitor needed in the patent application US 2010/0238689 is hard to realize in real world.
The present disclosure provides a precisely controlled constant current controller which is realized with simple circuit.